Signal processing method, signal processing device, and signal processing program

ABSTRACT

Provided is a signal processing method which reduces a plurality of echoes by receiving a plurality of reception signals and subtracting a pseudo echo generated by a plurality of adaptive filters which input the reception signals from a plurality of echoes generated by the reception signals. At least one of the reception signals is delayed to generate a delayed reception signal. The reception signal and the delayed reception signal are inputted to the adaptive filters to generate a pseudo echo. The frequency of inputting the reception signal and the delayed reception signal to the adaptive filters is controlled in accordance with the sensitivity of a localization change of the reception signals.

CROSS REFERENCE TO RELATED APPLICATION

This application is a National Stage of International Application No.PCT/JP2009/066068 filed Sep. 15, 2009, claiming priority based onJapanese Patent Application No. 2008-247273, filed Sep. 26, 2008, thecontents of all of which are incorporated herein by reference in theirentirety.

TECHNICAL FIELD

The present invention relates to a signal processing method, a signalprocessing apparatus, and a signal processing program.

BACKGROUND ART

As a signal processing apparatus for canceling echoes caused by receivedsignals propagating through spatial acoustic paths in a system usingplural received signals and a single or plural transmission signals, theapparatus of a linear coupled type is disclosed in Non-PatentLiterature 1. A block diagram of a multichannel signal processingapparatus of a linear coupled type in the case that the number ofchannels is two, namely, in the case of the signal processing apparatushaving a stereo signal as a target, is shown in FIG. 18. According toPatent Literature 1, the apparatus of the linear coupled type has aproblem that the coefficients have an indefinite number, namely aproblem that adaptive filter coefficients converge to indefinite valuesother than values equal to the features of the echo paths (correctsolutions). The filter coefficient value that has converged depends upona cross correlation between the filter input signals, and thus, when thecross correlation is changed due to movement etc. of a far-end talker,the correct coefficient value also varies. A variation in the correctcoefficient value caused by a change in the cross correlation means thatthe echo canceling capability is degraded even with no variation in theecho paths. Thus, the residual echoes are perceived, and thus, a speechquality is degraded.

In order to solve this problem, the signal processing method in which asingle adaptive filter per channel estimates echoes caused by signalspropagating from a single sound source in a plurality of paths bygenerating pseudo echoes (echo replicas) with adaptive filterscorresponding one to one to mixed signals with one received signal as aninput is disclosed in Patent Literature 1. A problem that thecoefficients have an indefinite number does not exist in this signalprocessing method because one adaptive filter cancels the echo to begenerated on one channel. As a result, the adaptive filter coefficientsconverge to optimum values that are uniquely determined. However, theNon-patent Literature 2 discloses evaluation results proving that thefact that the echo canceling capability is degraded when the parametersdetermined by the environment in use such as the locations ofmicrophones receiving the taker's voice are not within a certain range.Hence, in order to use the cancellation apparatus in a wide variety ofenvironments, a multichannel echo canceller of a linear coupled type hasto be used.

Based upon this premise, Patent Literature 2 discloses the methodcapable of uniquely determining the adaptive filter coefficients bydelaying the received signal of the multichannel echo canceller of thelinear coupled type, thereby to generate the delayed signal, andcontinuously and mutually alternating this as a new received signal withthe received signal. In this signal processing method, the number ofconditionals used to calculate the adaptive filter coefficients isincreased because of the introduction of the delayed received signals,whereby a problem that the solution, being the adaptive filtercoefficient, becomes indefinite does not occur. As a result, theadaptive filter coefficients converge to optimum values that areuniquely determined. However, with the case of the method proposed inthe Patent Literature 2, a movement of the acoustic image may often beperceived when the received signal and the delayed received signal areswitched. The movement of the acoustic image is perceived as anunnatural sound because it seems as if the acoustic image had moved eventhough it does not move as a matter of fact, and hence, a subjectivesound quality of the received signals is degraded. In order to solvethis, the method of correcting the amplitudes of signals in bothchannels when the received signal and the delayed received signal areswitched is disclosed in Patent Literature 3.

On the other hand, the method capable of uniquely determining theadaptive filter coefficients by applying a non-linear processing to thereceived signals in both channels instead of switching the receivedsignal and the delayed received signal is disclosed in Non-patentLiterature 3. However, Non-patent Literature 4 makes it clear that themethods disclosed in the Patent Literature 2, the Patent Literature 3,and the Non-patent Literature 3 provide a slow convergence rate,respectively, as compared with the multichannel echo canceller of thelinear coupled type. It is shown in the Non-patent Literature 4 that themethod disclosed in the Non-patent Literature 3 provides a yet slowerconvergence rate as compared with each of the method disclosed in thePatent Literature 2 and the method disclosed in the Patent Literature 3.

CITATION LIST Patent Literature

-   PTL 1: JP-P1992-284732A-   PTL 2: JP-P1999-004183A-   PTL 3: JP-P2000-078061A

Non-Patent Literature

-   NON-PTL 1: The Technical Report of the institute of Electronics,    Information and Communication Engineers (IEICE) of Japan, Vol. 84,    No. 330, pp. 7-14, CS-84-178-   NON-PTL 2: IEEE Proceedings of International Conference on    Acoustics, Speech and Signal Processing, Vol. 2, 1994, pp. 245-248-   NON-PTL 3: IEEE Proceedings of International Conference on    Acoustics, Speech and Signal Processing, Vol. 1, 1997, pp. 303-306-   NON-PTL 4: IEEE Proceedings of International Conference on    Acoustics, Speech and Signal Processing, Vol. 6, 1998, pp. 3677-3680

SUMMARY OF INVENTION Technical Problem

Each of the methods disclosed in the Patent Literature 3 and theNon-patent Literature 3 is slow in the convergence rate as compared withthe signal processing apparatus of the linear coupled type. Further, themethod disclosed in the Patent Literature 3 has a problem that makingthe convergence rate fast often causes a movement of the acoustic imagelocalization to be perceived, and hence the subjective sound quality ofthe received signals is degraded. Thus, the method disclosed in thePatent Literature 3 is not able to simultaneously accomplish a shortconvergence time and a high subjective sound quality.

Thereupon, the present invention has been accomplished in considerationof the above-mentioned problems, and an object thereof is to provide asignal processing method, a signal processing apparatus, and a signalprocessing program with an excellent subjective sound quality of thereceived signals and a short convergence time (a fast convergence rate),wherein the coefficient values of the adaptive filters converge tocorrect values that are uniquely determined by impulse responses of theecho paths.

Solution to Problem

The present invention is a signal processing method of receiving aplurality of received signals, and subtracting echo replicas generatedby a plurality of adaptive filters having said plurality of receivedsignals as an input, respectively, from a plurality of echoes to begenerated from said plurality of received signals, thereby to reducesaid plurality of echoes, comprising: generating delayed receivedsignals by delaying at least one received signal, out of said pluralityof received signals; generating echo replicas by inputting said receivedsignals and said delayed received signals into said adaptive filters;and controlling a frequency of inputting said received signals and saiddelayed received signals into said adaptive filters based upon aperceptual sensitivity to a change in acoustic image of said pluralityof received signals.

In additions, the present invention is a signal processing apparatus forreceiving a plurality of received signals, and subtracting echo replicasgenerated by a plurality of adaptive filters having said plurality ofreceived signals as an input, respectively, from a plurality of echoesto be generated from said plurality of received signals, thereby toreduce said plurality of echoes, comprising: a linear processing circuitfor generating delayed received signals by delaying at least onereceived signal, out of said plurality of received signals; an adaptivefilter for generating echo replicas by receiving said received signalsand said delayed received signals, a plurality of subtracters eachgenerating echo-reduced signals by subtracting said echo replicas from aplurality of mixed signals; and an analyzing circuit for obtaining aperceptual sensitivity to a change in acoustic image localization ofsaid plurality of received signals, and wherein said signal processingapparatus controlling a frequency of inputting said received signals andsaid delayed received signals into said adaptive filters based upon saidperceptual sensitivity, and controlling coefficients of said pluralityof adaptive filters so that outputs of said plurality of subtracters areminimized.

In additions, the present invention is a signal processing program forcausing a computer to execute a receiving process of receiving aplurality of received signals, and an echo reducing process of reducinga plurality of echoes that are generated by said plurality of receivedsignals, said signal processing program comprising: a delayed receivedsignal generating process of generating delayed received signals bydelaying at least one received signal, out of said plurality of receivedsignals; an echo replica generating process of generating echo replicasby inputting said received signals and said delayed received signalsinto said adaptive filters; and an echo replica subtracting process ofsubtracting said echo replicas from said plurality of received signals,respectively, wherein a frequency of inputting said received signals andsaid delayed received signals into said adaptive filters is controlledbased upon a perceptual sensitivity to a change in acoustic imagelocalization of said plurality of received signals.

Advantageous Effect of Invention

The signal processing method, the signal processing apparatus, and thesignal processing program of the present invention generate delayedreceived signals by delaying at least one received signal, and activatethe adaptive filters with the foregoing received signals and theforegoing delayed received signals taken as an input, respectively. Thenumber of conditionals at the moment of obtaining the adaptive filtercoefficients is increased because both of the received signal and thedelayed received signal are used, and thus, the problem that thesolutions become indefinite does not occur. Hence, the adaptive filtercoefficients converge to the optimum values that are uniquelydetermined.

Further, a frequency of inputting the foregoing received signals and theforegoing delayed received signals into the foregoing adaptive filtersis controlled based upon a perceptual sensitivity to a change in theacoustic image localization of the foregoing plurality of receivedsignals. This enables the foregoing received signals and the foregoingdelayed received signals to be inputted into the foregoing adaptivefilters according to a status of the signals at a frequency with whichthe subjective perception is avoided, and a degradation in thesubjective sound quality to be made small.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a block diagram illustrating a best mode of the signalprocessing apparatus of the present invention.

FIG. 2 is a block diagram illustrating an exemplary configuration of afilter 310.

FIG. 3 is a view illustrating an example of a variation in a coefficientc₀(k) in the filter 310.

FIG. 4 is block diagram illustrating a second exemplary configuration ofthe filters 310 and 320.

FIG. 5 is a view illustrating an example of a variation in thecoefficient c₀(k), a coefficient c₁(k), and a coefficient c₂(k) in thefilter 310.

FIG. 6 is block diagram illustrating a third exemplary configuration ofthe filter 310.

FIG. 7 is a block diagram illustrating a second embodiment of thepresent invention.

FIG. 8 is a view illustrating an example of a variation in a coefficientc₀(k) in the filters 310 and 320.

FIG. 9 is a view illustrating an example of a variation in thecoefficient c₀(k), the coefficient c₁(k), and the coefficient c₂(k) inthe filter 320.

FIG. 10 is a block diagram illustrating a third embodiment of thepresent invention.

FIG. 11 is a block diagram illustrating an exemplary configuration of anamplitude processing circuit 410.

FIG. 12 is a block diagram illustrating a fourth embodiment of thepresent invention.

FIG. 13 is a block diagram illustrating an exemplary configuration of anon-linear amplitude processing circuit 510.

FIG. 14 is a block diagram illustrating a fifth embodiment of thepresent invention.

FIG. 15 is a block diagram illustrating an exemplary configuration of alinear amplitude processing circuit 530.

FIG. 16 is a block diagram illustrating a sixth embodiment of thepresent invention.

FIG. 17 is a block diagram illustrating a seventh embodiment of thepresent invention.

FIG. 18 is a block diagram illustrating the signal processing apparatusof the linear coupled type.

DESCRIPTION OF EMBODIMENTS

The embodiments of the present invention will be explained in details byusing FIG. 1 to FIG. 18. It is now assumed to use a two-channel acousticecho canceller having a first received signal and a second receivedsignal that cancels the acoustic echoes caused by received signalspropagating from loudspeakers to microphones via spatial acoustic paths.

The signal processing apparatus of the present invention with tworeceived signals and two transmission signals, according to the bestmode, is shown in FIG. 1. A difference with the apparatus of the linearcoupled type disclosed in the Non-patent literature 1 lies in a pointthat the received signal 1 to be supplied to adaptive filters 122 and123 are processed by a delay processing circuit 300.

The received signals 1 and 2 are supplied to the delay processingcircuit 300. The delay processing circuit 300 generates the delayedreceived signals by delaying the received signal 1, and transmits themto adaptive filters 121 and 123 and a digital-analogue (DA) converter18, respectively. The delay processing circuit 300 transmits thereceived signal 2 to adaptive filters 122 and 124 and a digital-analogue(DA) converter 19, respectively. The DA converters 18 and 19 convert thedigital delayed received signals or the digital received signals intoanalogue delayed received signals or analogue received signals,respectively, and transmit them to loudspeakers 3 and 4, respectively.The loudspeakers 3 and 4, upon receipt of the delayed received signalsand the received signals, radiate them into the space. Microphones 9 and10, upon receipt of a talker 11's voice, and crosstalks (echoes) of thedelayed received signals radiated from the loudspeakers 3 and 4 into thespaces, transmits them to analogue-digital (AD) converters 20 and 21 asmixed signals 14 and 15, respectively. The AD converters 20 and 21convert the mixed signals 14 and 15 from the analogue signals into thedigital signals, and transmit them to subtracters 129 and 130. On theother hand, the adaptive filters 121 and 123 having received theforegoing delayed received signals and the adaptive filters 122 and 124having received the foregoing received signals generate the pseudoechoes (echo replicas), and transmit them to the subtracters 129 and130, respectively. The subtracters 129 and 130 subtract the echoreplicas generated by the adaptive filters 121 and 122, and the echoreplicas generated by the adaptive filters 123 and 124 from theforegoing mixed signals 14 and 15, respectively, and output them asoutput signals 16 and 17. With the subtraction by the subtracters 129and 130, the echoes are canceled, and as a result, the output signals 16and 17 are converted into the signals including no echo.

As algorithm for updating the coefficients of the adaptive filters 121,122, 123, and 124, the Least Mean Square (LMS) algorithm is disclosed inNon-patent Literature 4 (“Adaptive Signal Processing”, 1985, pp. 99-113,Prentice-Hall Inc., USA) and the Normalized Least Mean Square (NLMS)algorithm is disclosed in Non-patent Literature 5 (“Adaptive Filters”,1985, pp. 49-56, Kulwer Academic Publishers, USA).

As algorithm of the adaptive filter, the Sequential Regression Algorithm(SRA) described in the Non-patent Literature 4, and the RLS algorithmdescribed in the Non-patent Literature 5 may be also used.

The delayed signals are intermittently generated in the delay processingcircuit 300, and the signal obtained by delaying the received signal 1,and the signal equal to the received signal 1, that is, the not-delayedsignal coexist in the delayed received signal. The cross correlationbetween the signals to be supplied to the adaptive filters 121 and 123,and the signals to be supplied to the adaptive filters 122 and 124differs from each other in these two kinds of the statuses (in a statuswhere the received signals have been delayed and in a status where thereceived signals have not been delayed). For this, the two kinds of theconditionals (more than the conditionals of the apparatus of the linearcoupled type) responding to two kinds of the statuses can be gained, andthe coefficients of the adaptive filters 121, 123, 122, and 124 convergeto the correct values.

The relative delay amount (relative delay) of the delayed receivedsignal to the received signal can be set to an integer multiple of asampling period. In this case, the minimum value is equalized to thesampling period. The larger the amount of the relative delay, the largerthe cross correlation between the signals to be supplied to the adaptivefilters 121 and 123 and the signals to be supplied to the adaptivefilters 122 and 124, hence, the convergence time is shortened. However,the movement amount of the acoustic images at the moment that thedelayed received signals are reproduced in the loudspeakers 3 and 4becomes larger, and thus, the subjective sound quality is degraded. Forthis, the large relative delay is appropriately selected within anallowable range of the movement amount of the acoustic images that areperceived.

The relative delay can be also set to a non-integer multiple of thesampling period. In this case, the more suitable selection is enabledbecause a fine adjustment is enabled in a selection of the allowablemovement amount of the acoustic images and the relative delay that is aslarge as possible.

The relative delay does not need to be restricted to one kind, and aplurality of values can be also used alternately. For example, the threestatuses (two kinds of the relative delays), that is, a status in whichthe relative delay is a zero multiple of the sampling period (no delay),a status in which the relative delay is a one multiple of the samplingperiod (a one-sample delay), and a status in which the relative delay istwice as large as the sampling period (a two-sample delay) can beswitched. In this case, the number of the foregoing conditionals isthree times as large as that of the apparatus of the linear coupledtype, and hence, the foregoing adaptive filters can be caused toconverge in a far faster rate. As the number of the relative delayamounts that are utilized is increased, the foregoing adaptive filtersconverge at a faster rate.

The delay processing circuit 300 includes a filter 310, and an analyzingcircuit 350. The filter 310 generates the delayed received signal bydelaying the received signal 1. Further, the filter 310 outputs thereceived signal 1 as it stands without delaying it in some cases. Afrequency at which the output of the filter 310 varies between thedelayed received signal and the received signal 1 is controlled with aclock signal that is supplied from the analyzing circuit 350. Theadapter filterers 121 and 123, to which this switching signal issupplied, converge more quickly as this frequency becomes higher. Thereason is that with the switching, a cross correlation between thereceived signal 1 and the received signal 2 is changed, and hence acombination as well of the conditionals representing a relation betweenthe received signals 1 and 2, and the coefficients of the adaptivefilters 121 and 123 is changed. On the other hand, the switching allowsthe acoustic images that are localized by the loudspeakers 3 and 4 tomove toward the loudspeaker 4. Hence, a high switching frequency causesthe movement of the acoustic image localization to be easily perceivedsubjectively, which leads to a degradation in the subjective soundquality. That is, there is a trade-off between the convergence rate andthe subjective sound quality in terms of the foregoing switchingfrequency.

The analyzing circuit 350, upon receipt of the received signal 1 and thereceived signal 2, calculates the perceptual sensitivity to the movementof the acoustic images that are localized by these received signals. Theanalyzing circuit 350 generates the clock signal corresponding to theobtained perceptual sensitivity, and supplies it to the filter 310. Thefilter 310 decides a generating frequency of the delayed receivedsignals based upon the supplied clock signal.

A high perceptual sensitivity to the acoustic image movement by thereceived signal 1 and the received signal 2 means that the movement ofthe acoustic images is easily perceived. That is, even the slightmovement of the acoustic images is perceived. To the contrary, a lowperceptual sensitivity means that the movement of the acoustic images ishardly perceived. For example, when the received signal 1 and thereceived signal 2 resemble each other, it follows that the amplitudesand the phases of both resemble each other. In such a case, the acousticimages are localized in a position extremely close to the center betweenthe loudspeaker 3 for reproducing the received signal 1 and theloudspeaker 4 for reproducing the received signal 2. The movement of theacoustic images is easily perceived when the acoustic images are locatedmidway between the frontal loudspeakers 3 and 4. To the contrary, whenthe acoustic images are localized at a position far away from the centerof the loudspeakers 3 and 4, namely, are localized at a position closeto the loudspeaker 3 or the loudspeaker 4, it is not easy to perceivethe movement of the acoustic images. Thereupon, the analyzing circuit350 generates the clock signal in such a manner that a switchingfrequency of the received signal and the delayed receive signal becomeslow when the perceptual sensitivity is high, and a switching frequencyof the received signal and the delayed receive signal becomes high whenthe perceptual sensitivity is low, and supplies it to the filter 310.

An example of an index of the perceptual sensitivity, a similaritydegree that is typified by the cross correlation between the receivedsignal 1 and the received signal 2 may be used. The cross correlation,as a rule, can be expressed by a product of sample values at each timeof two signals. So as to obtain an averaged cross relation for anon-stationary signal, the quotient of the foregoing products divided bya total number of the samples accumulated for a constant time can bealso calculated. In addition, so as to avoid dependence of the obtainedquotient upon an absolute value of an input signal power, a normalizedcross correlation obtained by normalizing the foregoing accumulation ofthe products in terms of a product of a sum of squared sample values ofthe received signal 1 and a sum of squared sample values of the receivedsignal 2 each of which corresponds to an identical time can be alsoused. As a special version of the cross correlation, the phasecorrelation using the phases of the received signals and the amplitudecorrelation using the amplitudes of the received signals, an averagedvalue thereof, and a normalized value thereof can be also used. Inaddition, it is also possible that the analyzing circuit 350 isconfigured not to operate the correlation calculation when the signalpower or the amplitude is extremely low. Excluding small signal samplesthat are susceptible to an undesirable influence due to added noisemakes it possible to increase a tolerance to noise.

An example of another index of the perceptual sensitivity, powers of thereceived signal 1 and the received signal 2 may be used. When the powersof the received signal 1 and the received signal 2 are small, themovement of the acoustic images is hardly perceived even though theacoustic images are moved. The reason is that the small power makes itdifficult to listen to the received signals themselves. To the contrary,when the powers of the received signal 1 and the received signal 2 arelarge, the movement of the acoustic images is easily perceived. Thus,the analyzing circuit 350 generates the clock signal such that aswitching frequency between the received signal and the delayed receivesignal becomes low when the powers of the received signal 1 and thereceived signal 2 are large, and a switching frequency between thereceived signal and the delayed receive signal becomes high when thepowers of the received signal 1 and the received signal 2 are small, andsupplies it to the filter 310. Either one power can be used as an indexof the powers of the received signal 1 and received signal 2 because itcan be presumed that a correlation between both is high. Further, anaveraged value of the powers of the received signals 1 and 2 can be alsoused. In either case, not only an instantaneous value but also anaveraged value for a constant time can be also used. In addition, whenthe signal power or the amplitude is extremely low, it is also possiblethat the analyzing circuit 350 is configured not to operate thecorrelation calculation. Excluding small signal samples that aresusceptible to an undesirable influence due to added noise makes itpossible to increase a tolerance to noise.

FIG. 2 is a block diagram illustrating an exemplary configuration of thefilter 310. The filter is configured as a two-tap FIR filter having c₀and c₁ as the coefficient. The received signal 1 of FIG. 1 is suppliedto an input terminal 3100 of FIG. 2. The signal to be obtained in anoutput terminal 3104 of FIG. 2 is the delayed received signal.

The signal supplied to the input terminal 3100 is transmitted to a delayelement 3101 ₁ and a coefficient multiplier 3102 ₀. The coefficientmultiplier 3102 ₀ multiplies the inputted received signal sample by acoefficient value c₀ and transmits its product to an adder 3103 ₁. Thedelay element 3101 ₁ delays the received signal sample by one sample,and transmits it to a coefficient multiplier 3102 ₁.

The coefficient multiplier 3102 ₁ multiplies the inputted receivedsignal sample by a coefficient value c₁ and transmits its product to anadder 3103 ₁. The adder 3103 ₁ adds the output of the coefficientmultiplier 3102 ₀ and the output of the coefficient multiplier 3102_(k), and outputs its sum as the delayed received signal to the outputterminal 3104.

The clock signal, which is supplied to an input terminal 3105 from theanalyzing circuit 350 of FIG. 1, is transmitted to the coefficientmultiplier 3102 ₀, a coefficient multiplier 3102 ₁, and a coefficientmultiplier 3102 ₂. Based upon the clock signal supplied from the inputterminal 3105, each of the coefficient multiplier 3102 ₀, thecoefficient multiplier 3102 ₁, and the coefficient multiplier 3102 ₂varies its coefficient value.

The coefficient c₀ of the coefficient multiplier 3102 ₀ and thecoefficient c₁ of the coefficient multiplier 3102 ₁ vary with a time. Soas to clearly express this, c₀ and c₁ are denoted as c₀(k) and c₁(k),respectively. c₁(k) is given according to the following numericalequation.c ₁(k)=1−c ₀(k)  <Numerical equation 1>

One example of c₀(k) is shown in FIG. 3(A). i is assumed to be anarbitrary natural number. c₀(k) periodically has 1 and 0 every M(integer) samples. Further, as apparent from the numerical equation 1,c₁(k) varies as represented in the figure that is obtained by reversingFIG. 3(A) up and down. That is, c₀(k) and c₁(k) are exclusive to eachother, and either c₀(k) or c₁(k) is zero that is inputted into the adder3103. Hence, the output of the adder 3103 becomes equal to either thereceived signal or the delayed received signal, which is equivalent toswitching the received signal or the delayed received signal every Msamples. Additionally, while the maximum value of c₀(k) can be set to anarbitrary value, the output needs to be scaled by compensating a changein the amplitudes at that moment so that an output identical to theoutput that is gained when the maximum value of c₀(k) is 1 is yielded.

In FIG. 3(B) that differs from FIG. 3(A), c₀(k) is set so that it doesnot vary abruptly, but smoothly varies with a long lapse of a time atthe moment of varying between a zero value and a non-zero value. Asmooth variation in the value yields an effect that the acoustic imagesthat are generated at the moment of mutually switching the receivedsignal and the delayed received signal move smoothly, and the acousticimage movement is hardly perceived. Further, there is also an effect ofavoiding perception of the click sound at the moment of the foregoingswitching. This is effective in improving the subjective sound quality.

Upon comparing FIG. 3(B) with FIG. 3(C), the time of c₀(k)=1 and thetime of c₀(k)=0 differ from each other. The convergence of the adaptivefilter coefficients to the corrective values can be accomplished for ashortest time when each of the time of c₀(k)=1 and the time of c₀(k)=0becomes maximum because the cross correlation between the signal to besupplied to the adaptive filters 121 and 123 and the signal to besupplied to the adaptive filters 122 and 124 most largely differs fromthat of the apparatus of the linear coupled type when c₀(k)=0. In otherwords, the shorter the section in which the foregoing smooth variationin the value occurs, the shorter the convergence time. On the otherhand, it is felt that the movement of the acoustic images is abrupt allthe more as the section in which the foregoing smooth variation in thevalue occurs becomes shorter. Hence, the section in which the foregoingsmooth variation in the value occurs is set to have an appropriatelength by taking into consideration both of the perception of theacoustic image movement and the convergence time. While FIGS. 3(B) and(C) show an example where a variation of c₀(k) from c₀(k)=1 to c₀(k)=0(or the contrary hereto) is proportional to a time, an arbitrary smoothcurved line or straight line for connecting c₀(k)=1 and c₀(k)=0 can beused.

FIG. 4 is a block diagram illustrating a second exemplary configurationof the filter 310. The filter is configured as a three-tap FIR filterhaving c₀, c₁, and c₂ as the coefficient. The received signal 1 of FIG.1 is supplied to an input terminal 3100 of FIG. 4. The signal to beobtained in an output terminal 3104 of FIG. 4 is the delayed receivedsignal.

The signal supplied to the input terminal 3100 is transmitted to a delayelement 3101 ₁ and a coefficient multiplier 3102 ₀.

The coefficient multiplier 3102 ₀ multiplies the inputted receivedsignal sample by a coefficient value c₀ and transmits its product to anadder 3103 ₁. The delay element 3101 ₁ delays the received signal sampleby one sample, and transmits it to a coefficient multiplier 3102 ₁ and adelay element 3101 ₂.

The coefficient multiplier 3102 ₁ multiplies the output of the delayelement 3101 ₁ by a coefficient value c₁ and transmits its product to anadder 3103 ₁. The adder 3103 ₁ adds the output of the coefficientmultiplier 3102 ₀ and the output of the coefficient multiplier 3102_(k), and outputs its sum to an adder 3103 ₂. The delay element 3101 ₂delays the output of the delay element 3101 ₁ by one sample, andtransmits it to a coefficient multiplier 3102 ₂.

The coefficient multiplier 3102 ₂ multiplies the output of the delayelement 3101 ₂ by a coefficient value c₁ and transmits its product to anadder 3103 ₂. The adder 3103 ₂ adds the output of the adder 3103 ₁ andthe output of the coefficient multiplier 3102 ₂, and outputs its sum asthe delayed received signal to an output terminal 3104.

The clock signal, which is supplied to the input terminal 3105 from theanalyzing circuit 350 of FIG. 1, is transmitted to the coefficientmultiplier 3102 ₀, the coefficient multiplier 3102 ₁, and thecoefficient multiplier 3102 ₂. Based upon the clock signal supplied fromthe input terminal 3105, each of the coefficient multiplier 3102 ₀, thecoefficient multiplier 3102 ₁, and the coefficient multiplier 3102 ₂varies its coefficient value.

An example of the coefficient c₀(k) of the coefficient multiplier 3102₀, the coefficient c₁(k) of the coefficient multiplier 3102 ₁, and thecoefficient c₂(k) of the coefficient multiplier 3102 ₂ is shown in FIG.5. The coefficient c₀(k), the coefficient c₁(k), and the coefficientc₂(k) have 1 exclusively to each other, thereby allowing the receivedsignals subjected to the delay, which correspond to respectivecoefficient multipliers, to be gained as the delayed received signals inthe output terminal 3104. Like FIGS. 3(B) and (C) corresponding to FIG.3(A), c₀(k), c₁(k), and c₂(k) shown in FIG. 5 can be set so that theysmoothly vary at the moment of varying between a zero value and anon-zero value. A smooth variation in the value yields an effect thatthe acoustic images that are generated at the moment of mutuallyswitching the received signal and the delayed received signal movesmoothly, and the acoustic image movement is hardly perceived. Further,there is also an effect of avoiding perception of the click sound at themoment of the foregoing switching. This is effective in improving thesubjective sound quality.

FIG. 6 is a block diagram illustrating a third exemplary configurationof the filter 310. The filter is configured as an L-tap FIR filterhaving c₀, c₁, . . . , c_(L-1) as the coefficient. The received signal 1of FIG. 1 is supplied to an input terminal 3100 of FIG. 6. The signal tobe obtained in an output terminal 3104 of FIG. 6 is the delayed receivedsignal.

The signal supplied to the input terminal 3100 is transmitted to a delayelement 3101 ₁ and a coefficient multiplier 3102 ₀.

The coefficient multiplier 3102 ₀ multiplies the inputted receivedsignal sample by the coefficient value c₀ and transmits its product toan adder 3103 ₁. The delay element 3101 ₁ delays the received signalsample by one sample, and transmits it to a coefficient multiplier 3102₁ and a delay element 3101 ₂.

The coefficient multiplier 3102 ₁ multiplies the output of the delayelement 3101 ₁ by a coefficient value c₁ and transmits its product to anadder 3103 ₁. The adder 3103 ₁ adds the output of the coefficientmultiplier 3102 ₀ and the output of the coefficient multiplier 3102 ₁,and transmits its sum to an adder 3103 ₂. The delay element 3101 ₂delays the output of the delay element 3101 ₁ by one sample, andtransmits it to a coefficient multiplier 3102 ₂. Hereinafter, thisprocessing is repeated up to a coefficient multiplier 3102 _(L-2).

A coefficient multiplier 3102 _(L-1) multiplies the output of a delayelement 3101 _(L-1) by a coefficient value c_(L-1) and transmits itsproduct to an adder 3103 _(L-1). The adder 3103 _(L-1) adds the outputof an adder 3103 _(L-2) and the output of the coefficient multiplier3102 _(L-1), and outputs its sum as the delayed received signal to anoutput terminal 3104.

The clock signal, which is supplied to an input terminal 3105 from theanalyzing circuit 350 of FIG. 1, is transmitted to the coefficientmultiplier 3102 ₀, the coefficient multiplier 3102 ₁, . . . , and thecoefficient multiplier 3102 _(L-1). Based upon the clock signal suppliedfrom the input terminal 3105, each of the coefficient multiplier 3102 ₀,the coefficient multiplier 3102 ₁, . . . , and the coefficientmultiplier 3102 _(L-1) varies its coefficient value.

It may be considered that the coefficient c₀(k) of the coefficientmultiplier 3102 ₀, the coefficient c₁(k) of the coefficient multiplier3102 ₁, . . . , and the coefficient c_(L-1)(k) of the coefficientmultiplier 3102 _(L-1) correspond to respective taps of the filter 310connected in parallel. In other words, the coefficient c₀(k), thecoefficient c₁(k), . . . , and the coefficient c_(L-1)(k) have anon-zero value exclusively, and when one coefficient is non-zero, theother coefficients become zero. As explained by using FIG. 3(A) and FIG.5, c₀(k), c₁(k), . . . , and c_(L-1)(k) have non-zero exclusively toeach other, thereby allowing the received signals subjected to thedelay, which correspond to respective coefficient multipliers, to begained as the delayed received signals in the output terminal 3104. LikeFIGS. 3(B) and (C) corresponding to FIG. 3(A), c₀(k), c₁(k), . . . , andc_(L-1)(k) can be also set so that they smoothly vary at the moment ofvarying between a zero value and a non-zero value. A smooth variation inthe value yields an effect that the acoustic images that are generatedat the moment of mutually switching the received signal and the delayedreceived signal move smoothly, and the acoustic image movement is hardlyperceived. Further, there is also an effect of avoiding perception ofthe click sound at the moment of the foregoing switching. This iseffective in improving the subjective sound quality.

While the explanation was made so far on the assumption that each of thedelay amounts of the delay elements 3101 ₁, 3101 ₂, . . . , and 3101_(L-1) was equal to a one-sampling period, the delay amount may be aninteger multiple of the sampling period. Further, respective delayelements may be configured to give different delay amounts,respectively. Not limiting the delay amount of each delay element to theone-sampling period makes it possible to efficiently set the delays ofthe received signals to different various values, respectively.

Further, while the explanation was made so far on the assumption thatthe filter 310 had a configuration of the FIR filter, the filter mayhave the other structures such as a combination of a variable delaycircuit and a switch, and a combination of a variable delay circuit anda variable weighting mixing circuit so long as they have a configurationcapable of switching and outputting the received signal and delayedreceive signal with a time. Generating a plurality of the delayedreceived signals by giving different delays to the received signal witha plurality of the variable delay circuits, switching a plurality ofthese delayed received signals and received signal with the switch in acertain case, and appropriately mixing them with the variable weightingmixing circuit in another case makes it possible to realize a functionsimilar to that of a time-varying-coefficient FIR filter.

As explained above in details, the best mode of the present inventiongenerates the delayed received signals by delaying at least one receivedsignal, and activates the adaptive filters with the foregoing receivedsignals and the foregoing delayed received signals taken as an input,respectively. The number of conditionals at the moment of obtaining theadaptive filter coefficients is increased because both of the receivedsignal and the delayed received signal are used, and thus, the problemthat the solutions become indefinite does not occur. Hence, the adaptivefilter coefficients converge to the optimum values that are uniquelydetermined.

Further, a frequency of inputting the foregoing received signals and theforegoing delayed received signals into the foregoing adaptive filtersis controlled based upon a perceptual sensitivity to a change in theacoustic image localization of the foregoing plurality of receivedsignals. This enables the foregoing received signals and the foregoingdelayed received signals to be inputted into the foregoing adaptivefilters according to a status of the signals at a frequency with whichthe subjective perception is avoided, and a degradation in thesubjective sound quality to be made small.

The signal processing apparatus of the present invention with tworeceived signals and two transmission signals, according to the secondembodiment, is shown in FIG. 7. A difference with the best modeexplained by using FIG. 1 and FIG. 6 lies in a point of including adelay processing circuit 301 instead of the delay processing circuit300. Hereinafter, this difference will be explained in details.

The delay processing circuit 301 generates the delayed received signalsby delaying the received signal 1 and the received signal 2 andtransmits them to the adaptive filters 121 and 123 and thedigital-analogue (DA) converter 18 as well as the adaptive filters 122and 124 and the DA converter 19, respectively.

The delay processing circuit 301 includes filters 310 and 320 and aanalyzing circuit 351. The filters 310 and 320 generate the delayedreceived signals by delaying the received signal 1 and the receivedsignal 2. Further, the filters 310 and 320 output the received signal 1and the received signal 2 as they stand without delaying them in somecases. A frequency at which the output of the filter 310 varies betweenthe delayed received signal and the received signal 1 or the output ofthe filter 320 varies between the delayed received signal and thereceived signal 2 is controlled by the clock signal that is suppliedfrom the analyzing circuit 351. The adapter filterers 121 and 123 aswell as the adapter filterers 122 and 124, to which this switchingsignal is supplied, converge more quickly as this frequency becomeshigher. The reason is that a cross correlation between the receivedsignal 1 and the received signal 2 is changed by this switching, andhence a combination of the conditionals representing a relation betweenthe received signals 1 and 2, and the coefficients of the adaptivefilters 121 and 123 as well as a combination of the conditionalsrepresenting a relation between the received signals 1 and 2, and thecoefficients of the adaptive filters 122 and 124 are also changed. Onthe other hand, the acoustic images that are localized by theloudspeakers 3 and 4 move toward the loudspeaker 4 with the switching bythe filter 310 and move toward the loudspeaker 3 with the switching bythe filter 320. Hence, a high switching frequency causes the movement ofthe acoustic image localization to be easily perceived subjectively,which leads to a degradation in the subjective sound quality. That is,there is a trade-off between the convergence rate and the subjectivesound quality in terms of the foregoing switching frequency.

The analyzing circuit 351, upon receipt of the received signal 1 and thereceived signal 2, calculates the perceptual sensitivity to the movementof the acoustic images that are localized by these received signals. Theanalyzing circuit 351 generates the clock signals corresponding to theobtained perceptual sensitivity, and supplies them to the filters 310and 320, respectively. The filters 310 and 320 decide a generatingfrequency of the delayed received signals based upon the supplied clocksignals, respectively. Further, the shift of the phase of the clocksignal to be supplied to the filters 310 and that of the phase of theclock signal to be supplied to the filters 320 differ from each other.This phase shift will be explained later by using FIG. 8.

A configuration of the filer 320 is completely identical to that of thefilter 310 explained by using FIG. 2, FIG. 4, and FIG. 6. Further, thefilter 320, similarly to the filter 310, may have the other structuressuch as a combination of a variable delay circuit and a switch, and acombination of a variable delay circuit and a variable weighting mixingcircuit so long as they have a configuration capable of switching andoutputting the received signal and delayed receive signal with a time.Generating a plurality of the delayed received signals by givingdifferent delays to the received signal with a plurality of the variabledelay circuits, switching a plurality of these delayed received signalsand received signal with the switch in a certain case, and appropriatelymixing them with the variable weighting mixing circuit in another casemakes it possible to realize a function similar to that of atime-varying-coefficient FIR filter.

In FIG. 8, an example of a variation in c₀(k) in the filter 310 and thefilter 320 is shown on the assumption that the filters are configured asa two-tap FIR filter. When c₀(k) of the filter 310 is varied accordingto FIG. 8(A), c₀(k) of the filter 320 is varied according to FIG. 8(B).When c₀(k) of the filter 310 and the filter 320 is varied according toFIGS. 8(A) and (B), at least a moment that one outputs the receivedsignal and the other outputs the delayed received signal exists. In anexample of FIG. 8, the output of the filter 310 is the received signaland the output of the filter 320 is the delayed received signal at M/2sample just before 2iM. This status is defined as status 1. Further,contrarily, the output of the filter 310 is the delayed received signaland the output of the filter 320 is the received signal at M/2 samplejust before (2i+1)M. This status is defined as status 2. The outputs ofboth of the filters 310 and 320 are the received signals at M/2 samplejust after (2i−1)M, and the outputs of both of the filters 310 and 320are the delayed received signals at M/2 sample just after 2iM. Such astatus in which the output of the filter 310 and the output of thefilter 320 are identical to the received signals or the delayed receivedsignals all alike is defined as status 3. The cross correlation betweenthe signal to be supplied to the adaptive filters 121 and 123, and thesignal to be supplied to the adaptive filters 122 and 124 in the status3 is equal to that in the case of the apparatus of the linear coupledtype. That is, the cross correlation between the signal to be suppliedto the adaptive filters 121 and 123, and the signal to be supplied tothe adaptive filters 122 and 124 in the status 3 is equal to that in thecase of not utilizing the delayed received signal. Switching this statusand the status 1, and updating the adaptive filter coefficients so thattwo kinds of the cross correlation statuses are simultaneously satisfiedenables the adaptive filter coefficients to converge to the correctcoefficients. In addition, combining the status 2, and updating theadaptive filter coefficients so that “three kinds of the crosscorrelation statuses” of the status 1, the status 2, and the status 3are simultaneously satisfied enables the adaptive filter coefficients toconverge to the correct values at a faster rate as compared with thecase of utilizing two kinds of the cross correlation statuses.

Particularly, when the maximum value of the relative delay of the outputsignal of the filter 310 to the output signal of the filter 320 is equalto the maximum value of the relative delay of the output signal of thefilter 320 to the output signal of the filter 310, a shift amount of theacoustic image localization to the left caused by a switching to thedelayed received signal and a shift amount to the right are equalized toeach other, and the acoustic images are perceived as if the acousticimages had fluctuated left-right symmetrically with a time. For example,in the above-mentioned status 1 and status 2, the relative delay of theoutput signal of the filter 310 to the output signal of the filter 320,which is 1, and the relative delay of the output signal of the filter320 to the output signal of the filter 310, which is 1, are equal toeach other. A degradation in the subjective sound quality is smallerwith such a left-right symmetrical fluctuation in the acoustic imageslocalization as compared with the asymmetric movement to either the leftor the right because the left-right symmetrical fluctuation is perceivedas a blur of the acoustic images.

The phase of c₀(k) in FIG. 8(A) and that of c₀(k) in FIG. 8(B) differfrom each other by M/2 sample. This shift of the phase could be a valueother than M/2 sample. Adjusting this shift of the phase allows thetheoretical convergence time to become shortest when the above-mentionedthree kinds of the cross correlation statuses appear equally. Further, aperiod of a variation in c₀(k) does not need to be always equal to M/2sample, and an arbitrary value can be selected. The clock signal havingthis phase shift is generated by the analyzing circuit 351 as alreadyexplained.

With the case that the filter 320 is configured as a three-tap FIRfilter shown in FIG. 4, the coefficient c₀(k) of the coefficientmultiplier 3102 ₀, the coefficient c₁(k) of the coefficient multiplier3102 ₁, the coefficient c₂(k) of the coefficient multiplier 3102 ₂ ofthe filter 320 will be explained, in contrast to that of the filter 310.An example of the coefficient c₀(k) of the coefficient multiplier 3102₀, the coefficient c₁(k) of the coefficient multiplier 3102 ₁, and thecoefficient c₂(k) of the coefficient multiplier 3102 ₂ of the filter 320corresponding to FIG. 5, is shown in FIG. 9. With a relation betweenFIG. 5 and FIG. 9, similarly to a relation between FIG. 8(A) and FIG.8(B), the varying point (phase) of the corresponding coefficient valueis shifted. Appropriately setting this shift of the phases makes itpossible to change the convergence time. Further, as explained in anexample of the filter 310, the coefficient c₀(k) of the coefficientmultiplier 3102 ₀, the coefficient c₁(k) of the coefficient multiplier3102 ₁, and the coefficient c₂(k) of the coefficient multiplier 3102 ₂can be set so that a variation from non-zero to zero (or the contraryhereto) is proportional to a time in a certain case, and can be set sothat they have an arbitrary smooth curved line or straight line forconnecting non-zero and zero in another case.

Also with the case that the filter 320 is configured as an L-tap FIRfilter shown in FIG. 6, the filter 310 and the filter 320 differ fromeach other in the phase of c₀(k) as explained by using FIGS. 8(A) and(B), FIG. 5, and FIG. 9. Appropriately setting this shift of the phasesmakes it possible to change the convergence time. Further, as explainedin an example of the filter 310, the coefficient c₀(k) of thecoefficient multiplier 3102 ₀, the coefficient c₁(k) of the coefficientmultiplier 3102 _(k), and the coefficient c_(L-1)(k) of the coefficientmultiplier 3102 _(L-1) can be set so that a variation from non-zero tozero (or the contrary hereto) is proportional to a time in a certaincase, and can be set so that they have an arbitrary smooth curved lineor straight line for connecting non-zero and zero in another case.

In addition, similarly to the case of the two-tap FIR filter and thethree-tap FIR filter, the foregoing coefficient values can be controlledso that the maximum value of the relative delay of the output signal ofthe filter 310 to the output signal of the filter 320 is equal to themaximum value of the relative delay of the output signal of the filter320 to the output signal of the filter 310. This condition, moregenerally, makes it possible to express that the maximum values of therelative delays of the delayed signals to the received signals in thechannel that are reproduced by the left and right loudspeakers locatedremotest from a center are equalized to each other. This condition isequivalent to a difference of the maximum value of the relative delaybetween the left channel and the right channels being zero. When theforegoing left and right loudspeakers are located asymmetrically withrespect to the center, the foregoing difference of the maximum value ofthe relative delay must be zero in a status of taking into considerationa bias of the acoustic images due to its asymmetry.

As explained above in details, the second embodiment of the presentinvention generates the delayed received signals by delaying tworeceived signals or more, and activates the adaptive filters with theforegoing received signals and the foregoing delayed received signalstaken as an input, respectively. The number of conditionals at themoment of obtaining the adaptive filter coefficients is increasedbecause both of the received signal and the delayed received signal areused, and thus, the problem that the solutions become indefinite doesnot occur. Hence, the adaptive filter coefficients converge to theoptimum values that are uniquely determined.

Further, a frequency of inputting the foregoing received signals and theforegoing delayed received signals into the foregoing adaptive filtersis controlled based upon a perceptual sensitivity to a change in theacoustic image localization of the foregoing plurality of receivedsignals. This enables the foregoing received signals and the foregoingdelayed received signals to be inputted into the foregoing adaptivefilters according to a status of the signals at a frequency with whichthe subjective perception is avoided, and a degradation in thesubjective sound quality to be made small.

Further, using two delayed received signals or more makes it possible tofurthermore increase the number of the foregoing conditionals and toshorten the convergence time of the solutions to the optimum values. Inaddition, generating the delayed received signals so that a differencebetween the left and right channels of the maximum value of a relativedelay of the delayed signal in a channel, which is reproduced by each ofthe left and right loudspeakers located remotest from a center, to thereceived signal is zero in a status of taking into consideration a biasof the acoustic images due to the left-right asymmetry in thearrangement of the foregoing left and right loudspeakers enables theshift amounts of the acoustic image localization to the left and theright caused by the delayed received signals to be equalized with eachother, and a degradation in the subjective sound quality to be madesmall.

The signal processing apparatus of the present invention with tworeceived signals and two transmission signals, according to the thirdembodiment, is shown in FIG. 10. A difference with the second embodimentexplained by using FIG. 7 to FIG. 9 lies in a point that the outputsignals of a delay processing circuit 301 are processed by an amplitudecorrecting circuit 400, and then, supplied to the adaptive filters 121,123, 122, and 124.

The amplitude correcting circuit 400 generates amplitude-correcteddelayed received signals by correcting the amplitudes of the delayedreceived signals, being outputs of the delay processing circuit 301, andtransmits them to the adaptive filters 121 and 123 and thedigital-analogue (DA) converter 18 as well as the adaptive filters 122and 124 and the DA converter 19, respectively.

The amplitude correction of the delayed received signals in theamplitude correcting circuit 400 is performed when the output of thedelay processing circuit 301 is equal to the delayed received signalobtained by delaying the received signal 1 or the received signal 2. Theamplitude correction makes it possible to correct a correlation of theamplitude of the signal between a plurality of channels, and to cancelthe shift of the acoustic image localization generated at the moment ofusing the delayed received signals instead of the received signals. Itis also possible to equalize the total power after the correction to thetotal power before the correction by making the correction for allchannels at the moment of the amplitude correction. Maintaining thetotal power of all channels at a constant level can eliminate thesubjective feeling of disorder at the moment that theamplitude-corrected signal and the amplitude-not-corrected signal areswitched.

The amplitude correcting circuit 400 includes amplitude processingcircuits 410 and 420. The amplitude processing circuit 410 corrects theamplitude of the delayed received signal generated by delaying thereceived signal 1, thereby to generate the amplitude-corrected delayedreceived signal. The amplitude processing circuit 420 corrects theamplitude of the delayed received signal generated by delaying thereceived signal 2, thereby to generate the amplitude-corrected delayedreceived signal. Each of the amplitude processing circuits 410 and 420can assume a completely identical configuration. The clock signals aresupplied to the amplitude processing circuits 410 and 420 from theanalyzing circuit 351 that is included in the delay processing circuit301. These clock signals are used for applying the amplitude correctionaccording to the timing in which the delayed signals are generated inthe delay processing circuit 301.

FIG. 11 is a block diagram illustrating an exemplary configuration ofthe amplitude processing circuit 410. The amplitude processing circuit410 is configured as a multiplier 4101 having g₀ as a coefficient. Thedelayed received signal obtained by delaying the received signal 1 issupplied to an input terminal 4100 of FIG. 11. The multiplier 4101increase the signal supplied to an input terminal 4100 by a factor ofg₀, and transmits it to an output terminal 4104. The signal to beobtained in the output terminal 4104 of FIG. 11 is the signal obtainedby increasing the delayed received signal supplied to the input terminal4100 by a factor of g₀.

The amplitude processing circuit 420 can assume a configuration that isobtained by using g₁ instead of g₀ as the coefficient of the multiplier4101 in FIG. 11, being a block diagram illustrating an exemplaryconfiguration of the amplitude processing circuit 410. g₀ and g₁ have 1when the received signal 1 and the received signal 2 are supplied to theamplitude processing circuit 410 and the amplitude processing circuit420, respectively, and otherwise, have a value other than 1 (g₀-bar andg₁-bar). Such a value for compensating the shift of the acoustic imagelocalization caused by the delayed received signal is set to g₀-bar andg₁-bar. Further, the setting can be also made so that the total powerafter the correction is equalized to the total power before thecorrection. Maintaining the total power of all channels at a constantlevel can eliminate the subjective feeling of disorder at the momentthat the amplitude-corrected signal and the amplitude-not-correctedsignal are switched.

The amplitude processing circuit 410 and the amplitude processingcircuit 420 operate complementarily. That is, the movement of theacoustic images is corrected by g₀-bar and g₁-bar. The principle ofcorrecting the movement of the acoustic images caused by a change in thedelay amount by the amplitude correction is disclosed in Non-patentLiterature 6 (“Medical Research Council Special Report”, No. 166, 1932,pp. 1-32), Non-patent Literature 7 (“Journal of Acoustical Society ofAmerica”, Vol. 32, 1960, pp. 685-692), and Non-patent Literature 8(“Journal of Acoustical Society of America”, Vol. 94, 1993, pp. 98-110).

In an example of FIG. 10, when the acoustic images of the acousticsignals to be reproduced for a talker 11 by loudspeakers 3 and 4 movetoward the loudspeaker 3, in order to correcting this and to return theacoustic images to an original status, the amplitudes of the signals tobe radiated from the loudspeaker 4 in the acoustic space are increased,and simultaneously, the amplitudes of the signals to be radiated fromthe loudspeaker 3 in the acoustic space are decreased.

According to the Non-patent Literature 8, in order to move the acousticimage by the amplitude correction, with the total power of the receivedsignal 1 and the received signal 2 maintained constant, the followingnumerical equation 2 must hold between respective powers P₁ [dB] and P₂[dB].P ₁ +P ₂ =C  <Numerical equation 2>

where C is a positive constant. Hence, when the powers of the receivedsignal 1 and the received signal 2 before the amplitude correction areP₁-bar [dB] and P₂-bar [dB], respectively, the following numericalequation 3 must hold for the powers P₁ [dB] and P₂ [dB] of the receivedsignal 1 and the received signal 2 after the amplitude correction.P ₁ =P ₁-bar−ΔP/2P ₂ =P ₂-bar−ΔP/2  <Numerical equation 3>

where ΔP is a power correction amount. For this reason, the values ofthe coefficients g₀-bar and g₁-bar of the multiplier 4101 can bedetermined with the following numerical equation from the numericalequation 3.g ₀-bar=10^(−ΔPi/40)g ₁-bar=10^(ΔPi/40)  <Numerical equation 4>

where ΔP_(i) is a power compensation coefficient required to compensatethe received signals delayed by i samples.

Additionally, in the explanation made so far, according to FIG. 10, theconfiguration was explained of generating the delayed received signalsby processing the received signals with the delay processing circuit301, generating the amplitude-corrected delayed received signals bycorrecting the amplitudes of the delayed received signals with theamplitude correcting circuit 400, and supplying them to the adaptivefilters 121, 123, 122, and 124. On the other hand, it is also possibleto assume the configuration in which the order of the processing of thereceived signals is exchanged, namely the configuration of generatingthe amplitude-corrected received signals by correcting the amplitudes ofthe received signals with the amplitude correcting circuit 400,generating the amplitude-corrected delayed received signals byprocessing the amplitude-corrected received signals with the delayprocessing circuit 301, and supplying them to the adaptive filters 121,123, 122, and 124. Configurations and operations of the delay processingcircuit 301 and the amplitude correcting circuit 400 at that moment havebeen already explained, so its explanation is omitted herein.

As explained above in details, the third embodiment of the presentinvention generates the delayed received signals by delaying tworeceived signals or more, and activates the adaptive filters with theforegoing received signals and the foregoing delayed received signalstaken as an input, respectively. The number of conditionals at themoment of obtaining the adaptive filter coefficients is increasedbecause both of the received signal and the delayed received signal areused, and thus, the problem that the solutions become indefinite doesnot occur. Hence, the adaptive filter coefficients converge to theoptimum values that are uniquely determined.

Further, a frequency of inputting the foregoing received signals and theforegoing delayed received signals into the foregoing adaptive filtersis controlled based upon a perceptual sensitivity to a change in theacoustic image localization of the foregoing plurality of receivedsignals. This enables the foregoing received signals and the foregoingdelayed received signals to be inputted into the foregoing adaptivefilters according to a status of the signals at a frequency with whichthe subjective perception is avoided, and a degradation in thesubjective sound quality to be made small.

Further, using two delayed received signals or more makes it possible tofurthermore increase the number of the foregoing conditionals and toshorten the convergence time of the solutions to the optimum values. Inaddition, generating the delayed received signals so that a differencebetween the left and right channels of the maximum value of a relativedelay of the delayed signal in a channel, which is reproduced by each ofthe left and right loudspeakers located remotest from a center, to thereceived signal is zero in a status of taking into consideration a biasof the acoustic images due to the left-right asymmetry in thearrangement of the foregoing left and right loudspeakers enables theshift amounts of the acoustic image localization to the left and theright caused by the delayed received signals to be equalized with eachother, and a degradation in the subjective sound quality to be madesmall.

Further, a degradation in the sound quality of the audible receivedsignals directly supplied to the loudspeaker is suppressed so that theexcellent sound quality can be maintained because the acoustic imagemovement caused by the introduction of the delayed received signals isoffset by the process of correcting the amplitudes of the inputtedsignals.

The signal processing apparatus of the present invention with tworeceived signals and two transmission signals, according to the fourthembodiment, is shown in FIG. 12. A difference with the third embodimentexplained by using FIG. 10 and FIG. 11 lies in a point that the outputsignals of the amplitude correcting circuit 400 are processed by anon-linear processing circuit 500, and then supplied to the adaptivefilters 121, 123, 122, and 124.

The non-linear processing circuit 500 generates non-linearamplitude-corrected delayed received signals by non-linearly processingthe amplitude-corrected delayed received signals, being outputs of theamplitude correcting circuit 400, and transmits them to the adaptivefilters 121 and 123 and the digital-analogue (DA) converter 18 as wellas the adaptive filters 122 and 124 and the DA converter 19,respectively. The non-linear amplitude-corrected delayed receivedsignals are smaller in the cross correlation between a plurality of thechannels than the amplitude-corrected delayed received signals. Hence,the convergence of the adaptive filters 121, 123, 122, and 124 can bemade yet faster.

The non-linear processing circuit 500 includes non-linear amplitudeprocessing circuits 510 and 520. The non-linear amplitude processingcircuit 510 non-linearly processes the amplitude of theamplitude-corrected delayed received signal obtained by delaying thereceived signal 1 and then correcting the amplitude thereof, thereby togenerate the non-linear amplitude-corrected delayed received signal. Thenon-linear amplitude processing circuit 520 non-linearly processes theamplitude of the amplitude-corrected delayed received signal obtained bydelaying the received signal 2 and then correcting the amplitudethereof, thereby to generate the non-linear amplitude-corrected delayedreceived signal. Each of the non-linear amplitude processing circuits510 and 520 can assume a completely identical configuration.

FIG. 13 is a block diagram illustrating an exemplary configuration ofthe non-linear amplitude processing circuit 510. The non-linearamplitude processing circuit 510 is configured of a coefficientmultiplier 512, a polarity determining circuit 513, a multiplier 514,and an adder 515. The amplitude-corrected delayed received signals,being outputs of the amplitude correcting circuit 400 of FIG. 12, aresupplied to an input terminal 511. The amplitude-corrected delayedreceived signals are transmitted to the coefficient multiplier 512, thepolarity determining circuit 513, and the adder 515. The coefficientmultiplier 512 increases its input signal by a factor of a, and outputsit. The polarity determining circuit 513 outputs 1 when the polarity ofthe signal supplied to the input is positive, and outputs 0 when it isnegative. The multiplier 514, to which the output of the coefficientmultiplier 512 and the output of the polarity determining circuit 513are supplied, transmits a product of both to the adder 515. Theamplitude-corrected delayed received signals are supplied to anotherinput terminal of the adder 515 as they stand. That is, it follows thatthe output of the adder 515 for a signal sample x(k) in an inputterminal 511 is (1+α)x(k) when the polarity of the input signal ispositive, and is x(k) when it is negative. This signal becomes theoutput signal of the non-linear amplitude processing circuits 510. Thatis, the non-linear amplitude processing circuits 510 constitutes ahalf-wave rectifier circuit. The non-linear amplitude processingcircuits 520 can assume a configuration completely identical to that ofthe non-linear amplitude processing circuits 510.

Additionally, in the explanation made so far, according to FIG. 12, theconfiguration was explained of generating the delayed received signalsby processing the received signals with the delay processing circuit301, generating the amplitude-corrected delayed received signals bycorrecting the amplitudes of the delayed received signals with theamplitude correcting circuit 400, generating the non-linearamplitude-corrected delayed received signals by processing theamplitude-corrected delayed received signals with the non-linearamplitude processing circuit 500, and supplying them to the adaptivefilters 121, 123, 122, and 124. On the other hand, it is also possibleto assume the configuration in which the order of the processing of thereceived signals is exchanged, namely the configuration of, afterprocessing the received signals in the order of the amplitudecorrection, the delay, and the non-linear processing, or in the order ofthe non-linear processing, the delay, and the amplitude correction,supplying them to the adaptive filters 121, 123, 122, and 124.Configurations and operations of the delay processing circuit 301, theamplitude correcting circuit 400, and the non-linear processing circuit500 at that moment have been already explained, so its explanation isomitted herein.

As explained above in details, the fourth embodiment of the presentinvention generates the delayed received signals by delaying tworeceived signals or more, generates the amplitude-corrected delayedreceived signals by correcting the amplitudes of the delayed receivedsignals, generates the non-linear amplitude-corrected delayed receivedsignals by non-linearly processing the amplitude-corrected delayedreceived signals, and activates the adaptive filters with the foregoingreceived signals and the foregoing non-linear amplitude-correcteddelayed received signals taken as an input, respectively. The number ofconditionals at the moment of obtaining the adaptive filter coefficientsis increased because both of the received signal and the non-linearamplitude-corrected delayed received signal are used, and thus, theproblem that the solutions become indefinite does not occur. Hence, theadaptive filter coefficients converge to the optimum values that areuniquely determined.

Further, a frequency of inputting the foregoing received signals and theforegoing delayed received signals into the foregoing adaptive filtersis controlled based upon a perceptual sensitivity to a change in theacoustic image localization of the foregoing plurality of receivedsignals. This enables the foregoing received signals and the foregoingdelayed received signals to be inputted into the foregoing adaptivefilters according to a status of the signals at a frequency with whichthe subjective perception is avoided, and a degradation in thesubjective sound quality to be made small.

Further, using a plurality of the delayed received signals makes itpossible to furthermore increase the number of the foregoingconditionals and to shorten the convergence time of the solutions to theoptimum values. In addition, generating the delayed received signals sothat a difference between the left and right channels of the maximumvalue of a relative delay of the delayed signal in a channel, which isreproduced by each of the left and the right loudspeakers locatedremotest from a center, to the received signal, is zero in a status oftaking into consideration a bias of the acoustic images due to theleft-right asymmetry in the arrangement of the foregoing left and rightloudspeakers enables the shift amounts of the acoustic imagelocalization to the left and the right caused by the delayed receivedsignals to be equalized with each other, and a degradation in thesubjective sound quality to be made small.

Further, a degradation in the sound quality of the audible receivedsignals directly supplied to the loudspeaker is suppressed so that theexcellent sound quality can be maintained because the acoustic imagemovement caused by the introduction of the delayed received signals isoffset by the process of correcting the amplitudes of the inputtedsignals. In addition, the convergence time can be shortened all the moreby a synergistic effect of the non-linear processing and theintroduction of the delayed received signals.

The signal processing apparatus of the present invention with tworeceived signals and two transmission signals, according to the fifthembodiment, is shown in FIG. 14. A difference with the fourth embodimentexplained by using FIG. 12 and FIG. 13 lies in a point that thenon-linear processing circuit 500 is replaced with a non-linearprocessing circuit 501.

The non-linear processing circuit 501 includes non-linear amplitudeprocessing circuits 530 and 540. The non-linear amplitude processingcircuit 530 non-linearly processes the amplitude-corrected delayedreceived signal obtained by delaying the received signal 1 and thencorrecting the amplitude thereof by using the amplitude-correcteddelayed received signal obtained by delaying the received signal 2 andthen correcting the amplitude thereof, thereby to generate thenon-linear amplitude-corrected delayed received signals. The non-linearamplitude processing circuit 540 non-linearly processes theamplitude-corrected delayed received signal obtained by delaying thereceived signal 2 and then correcting the amplitude thereof by using theamplitude-corrected delayed received signal obtained by delaying thereceived signal 1 and then correcting the amplitude thereof, thereby togenerate the non-linear amplitude-corrected delayed received signals.Each of the non-linear amplitude processing circuits 530 and 540 canassume a completely identical configuration.

FIG. 15 is a block diagram illustrating an exemplary configuration ofthe non-linear amplitude processing circuit 530. The non-linearamplitude processing circuit 530 is configured of a coefficientmultiplier 512, a polarity determining circuit 513, a multiplier 514,and an adder 515. The amplitude-corrected delayed received signalobtained by delaying the received signal 1 and then correcting theamplitude thereof, out of the outputs of the amplitude correctingcircuit 400 of FIG. 14, is supplied to an input terminal 531. Theamplitude-corrected delayed received signal obtained by delaying thereceived signal 2 and then correcting the amplitude thereof, out of theoutputs of the amplitude correcting circuit 400 of FIG. 14, is suppliedto an input terminal 537. The amplitude-corrected delayed receivedsignal generated from the received signal 1 is transmitted to thepolarity determining circuit 513 and the adder 515. Theamplitude-corrected delayed received signal generated from the receivedsignal 2 is transmitted to the coefficient multiplier 512. Thecoefficient multiplier 512 increases its input signal by a factor of a,and outputs it. The polarity determining circuit 513 outputs 1 when thepolarity of the signal supplied to the input is positive, and outputs 0when it is negative. The multiplier 514, to which the output of thecoefficient multiplier 512 and the output of the polarity determiningcircuit 513 are supplied, transmits a product of both to the adder 515.The amplitude-corrected delayed received signal generated from thereceived signal 1 is supplied to another input terminal of the adder 15as it stands. That is, it follows that the output of the adder 515 for asignal sample x₁(k) in an input terminal 531 and for a signal samplex₂(k) in an input terminal 537 is x₁(k)+αx₂(k) when the polarity of theinput signal is positive, and is x₁(k) when it is negative. This signalbecomes the output signal of the non-linear amplitude processingcircuits 530. The non-linear amplitude processing circuits 530 has theconfiguration in which the input of the coefficient multiplier 512 inthe non-linear amplitude processing circuits 510 has been changed fromthe amplitude-corrected delayed received signal generated from thereceived signal 1 to the amplitude-corrected delayed received signalgenerated from the received signal 2. The non-linear amplitudeprocessing circuits 540 can assume a configuration completely identicalto that of the non-linear amplitude processing circuits 530. In thisconfiguration, the variation from the signal not subjected to thenon-linear processing becomes large and an effect of reducing thecorrelation between the channels becomes large because the signalgenerated from the received signal of another channel is used for thenon-linear processing.

Additionally, in the explanation made so far, according to FIG. 14, theconfiguration was explained of generating the delayed received signalsby processing the received signals with the delay processing circuit301, generating the amplitude-corrected delayed received signals bycorrecting the amplitudes of the delayed received signals with theamplitude correcting circuit 400, generating the non-linearamplitude-corrected delayed received signals by processing theamplitude-corrected delayed received signals with the non-linearamplitude processing circuit 501, and supplying them to the adaptivefilters 121, 123, 122, and 124. On the other hand, it is also possibleto assume the configuration in which the order of the processing of thereceived signals is exchanged, namely the configuration of, afterprocessing the received signals in the order of the amplitudecorrection, the delay, and the non-linear processing, or in the order ofthe non-linear processing, the delay, and the amplitude correction,supplying them to the adaptive filters 121, 123, 122, and 124.Configurations and operations of the delay processing circuit 301, theamplitude correcting circuit 400, and the non-linear processing circuit501 at that moment have been already explained, so its explanation isomitted herein.

As explained above in details, the fifth embodiment of the presentinvention generates the delayed received signals by delaying tworeceived signals or more, generates the amplitude-corrected delayedreceived signals by correcting the amplitudes of the delayed receivedsignals, generates the non-linear amplitude-corrected delayed receivedsignals by non-linearly processing the amplitude-corrected delayedreceived signals, and activates the adaptive filters with the foregoingreceived signals and the foregoing non-linear amplitude-correcteddelayed received signals taken as an input, respectively. The number ofconditionals at the moment of obtaining the adaptive filter coefficientsis increased because both of the received signal and the non-linearamplitude-corrected delayed received signal are used, and thus, theproblem that the solutions become indefinite does not occur. Hence, theadaptive filter coefficients converge to the optimum values that areuniquely determined.

Further, a frequency of inputting the foregoing received signals and theforegoing delayed received signals into the foregoing adaptive filtersis controlled based upon a perceptual sensitivity to a change in theacoustic image localization of the foregoing plurality of receivedsignals. This enables the foregoing received signals and the foregoingdelayed received signals to be inputted into the foregoing adaptivefilters according to a status of the signals at a frequency with whichthe subjective perception is avoided, and a degradation in thesubjective sound quality to be made small.

Further, using a plurality of the delayed received signals makes itpossible to furthermore increase the number of the foregoingconditionals and to shorten the convergence time of the solutions to theoptimum values. In addition, generating the delayed received signals sothat a difference between the left and right channels of the maximumvalue of a relative delay of the delayed signal in a channel, which isreproduced by each of the left and right loudspeakers located remotestfrom a center, to the received signal is zero in a status of taking intoconsideration a bias of the acoustic images due to the left-rightasymmetry in the arrangement of the foregoing left and rightloudspeakers enables the shift amounts of the acoustic imagelocalization to the left and the right caused by the delayed receivedsignals to be equalized with each other, and a gradation in thesubjective sound quality to be made small. Further, a degradation in thesound quality of the audible received signals directly supplied to theloudspeaker is suppressed so that the excellent sound quality can bemaintained because the acoustic image movement caused by theintroduction of the delayed received signals is offset by the process ofcorrecting the amplitudes of the inputted signals. In addition, theconvergence time can be shortened all the more by a synergistic effectof the non-linear processing using the received signals of a pluralityof channels and the introduction of the delayed received signals.

The signal processing apparatus of the present invention with tworeceived signals and two transmission signals, according to the sixthembodiment, is shown in FIG. 16. A difference with the second embodimentexplained by using FIG. 7 to FIG. 9 lies in a point that a frequencyanalysis synthesizing circuit 600 is provided upstream of the delayprocessing circuit 301, and that a frequency analysis synthesizingcircuit 610 is provided upstream of the DA converters 18 and 19 as wellas downstream of the AD converters 20 and 21. Hence, all of the delayprocessing circuit 301, the adapter filters 121, 122, 123, and 124, andthe subtracters 129 and 130 are operative in response to band-dividednarrow band signals. The frequency analysis synthesizing circuit 600band-divided the received signals 1 and 2, and transmits them to thedelay processing circuit 301. The frequency analysis synthesizingcircuit 600 also band-synthesizes the outputs of the subtracters 129 and130, and constitutes all-band output signals 16 and 17. The frequencyanalysis synthesizing circuit 610 band-synthesizes the outputs of thedelay processing circuit 301, and transmits them to the DA converters 18and 19. The frequency analysis synthesizing circuit 610 alsoband-divides the outputs of the AD converters 20 and 21, and transmitsthem to the subtracters 129 and 130. The delay processing circuit 301adds the delays to the band-divided signals, and outputs them asband-divided delayed received signals. The sixth embodiment enables theoptimum delays to be given to the band-divided signals, respectively.Hence, it leads to an increase in a degree of freedom at the moment ofselecting the relative delay that is as large as possible within theallowable movement amount of the acoustic images, and an improvement inthe subjective sound quality, which was explained by using FIG. 1.

The frequency analysis function of the frequency analysis synthesizingcircuits 600 and 610 can be realized by applying a frequency conversionfor the input signal sample divided into the frames. As an example ofthe frequency conversion, a Fourier transform, a cosine transform, a KL(Karhunen Loeve) transform, etc. are known. The technology related to aspecific arithmetic operation of these transforms, and its propertiesare disclosed in Non-patent Literature 9 (DIGITAL CODING OF WAVEFORMS,PRINCIPLES AND APPLICATIONS TO SPEECH AND VIDEO, PRENTICE-HALL, 1990).Further, it is publicly known that other conversions such as a Hadamardtransform, a Haar transform, and a wavelet transform can be used.

The foregoing frequency analysis function can be also realized byapplying the foregoing transforms for a result obtained by weighting theinput signal samples of the above frame with a window function W. Assuch a window function, the window functions such as a Hamming window, aHanning (Hann) window, a Kaiser window, and a Blackman window are known.Further, more complicated window functions also can be used. Thetechnology related to these window functions is disclosed in Non-patentLiterature 10 and Non-patent Literature 11. In addition, the windowingas well by partially overlapping two continuous frames or more is widelycarried out. In this case, the foregoing frequency transforms are usedfor the signal subjected to the overlap windowing. The technologyrelated to the blocking involving the overlap and the conversion isdisclosed in the Non-patent Literature 10 (DIGITAL SIGNAL PROCESSING,PRENTICE-HALL, 1975).

In addition, the frequency analysis function of the frequency analysissynthesizing circuits 600 and 610 may be configured of a band-divisionfilter bank. The band-division filter bank is configured of a pluralityof band-pass filters. An interval of each frequency band of theband-division filter bank could be equal in a certain case, and unequalin another case. Carrying out the band division at an unequal intervalmakes it possible to lower/raise a time resolution, that is, the timeresolution can be lowered by carrying out the division into narrowsbands with regard to a low-frequency area, and the time resolution canbe raised by carrying out the division into wide bands with regard to ahigh-frequency area. As a typified example of the unequal-intervaldivision, there exists an octave division in which the band graduallyhalves toward the low-frequency area, a critical band division thatcorresponds to an auditory feature of a human being, or the like. Afterdividing into the frequency bands having an equal interval, a hybridfilter bank may be used for furthermore carrying out the band divisiononly with regard to a low-frequency area in order to enhance thefrequency resolution of the frequency bands in a low-frequency area. Thetechnology related to the band-division filter bank and its designmethod is disclosed in the Non-patent Literature 11 (MULTIRATE SYSTEMSAND FILTER BANKS, PRENTICE-HALL, 1993).

The frequency synthesis function of the frequency analysis synthesizingcircuits 600 and 610 is configured of an inverse conversioncorresponding to the frequency conversion for realizing the frequencyanalysis function of the frequency analysis synthesizing circuits 600and 610. When the frequency analysis function of the frequency analysissynthesizing circuits 600 and 610 includes the weighting by a windowfunction W, the frequency-synthesized signals are multiplied by thewindow function W. When the frequency analysis function of the frequencyanalysis synthesizing circuits 600 and 610 is configured of theband-division filter bank, the frequency synthesis function of thefrequency analysis synthesizing circuits 600 and 610 is configured of aband-synthesis filter bank. The technology related to the band-synthesisfilter bank and its design method is disclosed in the Non-patentLiterature 11.

Additionally, it is self-evident that a processing similar to theprocessing so far explained can be performed for the band-dividedsignals by combining the frequency analysis synthesizing circuits 600and 610, and any of the best mode and the third embodiment to the fifthembodiment of the present invention.

As explained above in details, the sixth embodiment of the presentinvention generates the band-divided received signals byfrequency-analyzing two received signals or more, generates theband-divided delayed received signals by delaying the above band-dividedreceived signals, and activates the adaptive filters with the foregoingband-divided received signals and the foregoing band-divided delayedreceived signals taken as an input, respectively. The number ofconditionals at the moment of obtaining the adaptive filter coefficientsis increased because both of the band-divided received signal and theband-divided delayed received signal are used, and thus, the problemthat the solutions become indefinite does not occur. Hence, the adaptivefilter coefficients converge to the optimum values that are uniquelydetermined.

Further, a frequency of inputting the foregoing received signals and theforegoing delayed received signals into the foregoing adaptive filtersis controlled based upon a perceptual sensitivity to a change in theacoustic image localization of the foregoing plurality of receivedsignals. This enables the foregoing received signals and the foregoingdelayed received signals to be inputted into the foregoing adaptivefilters according to a status of the signals at a frequency with whichthe subjective perception is avoided, and a degradation in thesubjective sound quality to be made small.

Further, using a plurality of the delayed received signals makes itpossible to furthermore increase the number of the foregoingconditionals and to shorten the convergence time of the solutions to theoptimum values. In addition, generating the delayed received signals sothat a difference between the left and right channels of the maximumvalue of a relative delay of the delayed signal in a channel, which isreproduced by each of the left and right loudspeakers located remotestfrom a center, to the received signal is zero in a status of taking intoconsideration a bias of the acoustic images due to the left-rightasymmetry in the arrangement of the foregoing left and rightloudspeakers enables the shift amounts of the acoustic imagelocalization to the left and the right caused by the delayed receivedsignals to be equalized with each other, and a degradation in thesubjective sound quality to be made small.

Further, the sixth embodiment enables the optimum delays to be given tothe band-divided signals, respectively, which leads to an increase in adegree of freedom at the moment of selecting the relative delay that isas large as possible within the allowable movement amount of theacoustic images, and an improvement in the subjective sound quality.

In the best mode and the second to sixth embodiments above, while theecho cancellation, with a multi-channel teleconference system as atarget, was discussed, a similar discussion holds also in asingle-channel multipoint teleconferencing system, being anotherapplication field of the signal processing. Normally, the single-channelmultipoint teleconferencing system performs the process of suitablyadding an attenuation and a time delay to the voice of a talker receivedby a single microphone such that the talker is localized in a desiredposition between a plurality of loudspeakers to be used at the receiveside. The signals processed in such a manner, of which the number isequivalent to that of the number of the loudspeakers at the receiveside, are generated. When the number of the loudspeakers at the receiveside is equal to 2, the two kinds of the signals, to which theabove-mentioned attenuation and delay have been added in the embodimentsshown in FIG. 1, FIG. 7, FIG. 10, FIG. 12, FIG. 14, and FIG. 16,correspond to the first received signal 1 and the second received signal2. Hence, the embodiments of the present invention can be appliedwithout any change.

While the case of using the first received signal 1 and the secondreceived signal 2, and the first mixed signal 14 and the second mixedsignal 15 was exemplified herein for explanation, the present inventionis applicable to the general case that plural received signals and asingle or plural transmission signals exist. Further, even though thedescription was performed with an example of the acoustic echoes inwhich the received signal propagated from the loudspeakers to themicrophone via the spatial acoustic paths and the acoustic echoesreceived by the microphone were cancelled, the present invention isapplicable to an application for canceling the echoes other than theacoustic echoes, for example, the echoes caused by the crosstalk etc. ina transmission line. Infinite impulse response adaptive filters may beused instead of the finite impulse response adaptive filters. Further,subband adaptive filters or transform-domain adaptive filters may beused.

Continuously, the seventh embodiment of the present invention will beexplained in details by making a reference to FIG. 17. The seventhembodiment of the present invention includes a computer 1000 that isoperative under a program control. The computer 1000 is operative basedupon a program for performing the processing related to any of theabove-mentioned best mode and second embodiment to sixth embodiment forthe received signals received from input terminals 1 and 2, andoutputting the signals of which the echoes have been canceled as outputsignals 16 and 17.

The first example is characterized in that a signal processing method ofreceiving a plurality of received signals, and subtracting echo replicasgenerated by a plurality of adaptive filters having said plurality ofreceived signals as an input, respectively, from a plurality of echoesto be generated from said plurality of received signals, thereby toreduce said plurality of echoes, comprising: generating delayed receivedsignals by delaying at least one received signal, out of said pluralityof received signals; generating echo replicas by inputting said receivedsignals and said delayed received signals into said adaptive filters;and controlling a frequency of inputting said received signals and saiddelayed received signals into said adaptive filters based upon aperceptual sensitivity to a change in 1 of received signals.

The second example in the above-mentioned example is characterized inthat at least one of said delayed received signals is anamplitude-corrected delayed received signal subjected to an amplitudecorrection.

The third example in the above-mentioned examples is characterized inthat at least one of said input signals of adaptive filters is anon-linearly processed signal subjected to a non-linear processing.

The fourth example in the above-mentioned examples is characterized inthat the signal processing method comprising decomposing said receivedsignal into a plurality of frequency components, and generating thedelayed received signals by delaying the received signal for each ofsaid plurality of frequency components.

The fifth example in the above-mentioned examples is characterized inthat the perceptual sensitivity to a change in acoustic imagelocalization is obtained based upon similarities of the receivedsignals.

The sixth example in the above-mentioned examples is characterized inthat the perceptual sensitivity to a change in acoustic imagelocalization is obtained based upon a power of the received signal.

The seventh example in the above-mentioned examples is characterized inthat the signal processing method comprising generating the delayedreceived signals whose relative delays to those of the originalundelayed received signals take different values and vary with time.

The eighth example in the above-mentioned examples is characterized inthat the relative delay is an integer multiple of a sampling period.

The ninth example in the above-mentioned examples is characterized inthat the delayed received signals are generated by processing thereceived signals with a filter having a plurality of time-varyingcoefficients which alternately take a zero or a non-zero value.

The tenth example in the above-mentioned examples is characterized thatthe time-varying coefficients have a zero value exclusively to eachother.

The eleventh example in the above-mentioned examples is characterized inthat the time-varying coefficients have a non-zero value exclusively toeach other.

The twelfth example is characterized in that a signal processingapparatus for receiving a plurality of received signals, and subtractingecho replicas generated by a plurality of adaptive filters having saidplurality of received signals as an input, respectively, from aplurality of echoes to be generated from said plurality of receivedsignals, thereby to reduce said plurality of echoes, comprising: alinear processing circuit for generating delayed received signals bydelaying at least one received signal, out of said plurality of receivedsignals; an adaptive filter for generating echo replicas by receivingsaid received signals and said delayed received signals, a plurality ofsubtracters each generating echo-reduced signals by subtracting saidecho replicas from a plurality of mixed signals; and an analyzingcircuit for obtaining a perceptual sensitivity to a change in acousticimage localization of said plurality of received signals, and whereinsaid signal processing apparatus controlling a frequency of inputtingsaid received signals and said delayed received signals into saidadaptive filters based upon said perceptual sensitivity, and controllingcoefficients of said plurality of adaptive filters so that outputs ofsaid plurality of subtracters are minimized.

The thirteenth example in the above-mentioned example is characterizedin that the signal processing apparatus comprising an amplitudecorrecting circuit for generating amplitude-corrected delayed receivedsignals by amplitude-correcting at least one signal, out of said delayedreceived signals.

The fourteenth example in the above-mentioned examples is characterizedin that the signal processing apparatus comprising a non-linearprocessing circuit for generating non-linearly processed signals bynon-linearly processing at least one signal, out of the signals to beinputted into said plurality of adaptive filters.

The fifteenth example in the above-mentioned examples is characterizedin that the signal processing apparatus comprising: a frequencyanalyzing circuit for decomposing said received signal into a pluralityof frequency components; and a linear processing circuit for generatingthe delayed received signals by delaying the received signal for everysaid plurality of frequency components.

The sixteenth example in the above-mentioned examples is characterizedin that the analyzing circuit obtains said perceptual sensitivity to achange in acoustic image localization based upon similarities of thereceived signals.

The seventeenth example in the above-mentioned examples is characterizedin that the analyzing circuit obtains said perceptual sensitivity to achange in localization based upon a power of the received signal.

The eighteenth example in the above-mentioned examples is characterizedin that the linear processing circuit performs a processing such thatrelative delays of said delayed received signals to said receivedsignals have a plurality of values that vary with a time.

The nineteenth example in the above-mentioned examples is characterizedin that the linear processing circuit performs a processing such thatsaid relative delay is an integer multiple of a sampling period.

The twentieth example in the above-mentioned examples is characterizedin that the linear processing circuit comprises a filter having aplurality of time-varying coefficients which alternately take a zero ora non-zero value.

The twenty-first example in the above-mentioned examples ischaracterized in that the time-varying coefficients have a zero valueexclusively to each other.

The twenty-second example in the above-mentioned examples ischaracterized in that the time-varying coefficients have a non-zerovalue exclusively to each other.

The twenty-third example is characterized in that a signal processingprogram for causing a computer to execute a receiving process ofreceiving a plurality of received signals, and an echo reducing processof reducing a plurality of echoes that are generated by said pluralityof received signals, said signal processing program comprising: adelayed received signal generating process of generating delayedreceived signals by delaying at least one received signal, out of saidplurality of received signals; an echo replica generating process ofgenerating echo replicas by inputting said received signals and saiddelayed received signals into said adaptive filters; and an echo replicasubtracting process of subtracting said echo replicas from saidplurality of received signals, respectively, wherein a frequency ofinputting said received signals and said delayed received signals intosaid adaptive filters is controlled based upon a perceptual sensitivityto a change in of received signals.

The twenty-fourth example in the above-mentioned example ischaracterized in that at least one of said delayed received signals isan amplitude-corrected delayed received signal subjected to an amplitudecorrection.

The twenty-fifth example in the above-mentioned examples ischaracterized in that at least one of said input signals of adaptivefilters is a non-linearly processed signal subjected to a non-linearprocessing.

The twenty-sixth example in the above-mentioned examples ischaracterized in that the signal processing program comprisingdecomposing said received signal into a plurality of frequencycomponents and generating the delayed received signals by delaying thereceived signal for every above plurality of frequency components.

The twenty-seventh example in the above-mentioned examples ischaracterized in that the perceptual sensitivity to a change inlocalization is obtained based upon similarities of the receivedsignals.

The twenty-eight example in the above-mentioned examples ischaracterized in that the perceptual sensitivity to a change inlocalization is obtained based upon a power of the received signal.

The twenty-nine example in the above-mentioned examples is characterizedin that the delayed received signals are generated so that relativedelays of said delayed received signals to said received delays have aplurality of values that vary with a time.

The thirty example in the above-mentioned examples is characterized inthat the relative delay is an integer multiple of a sampling period.

The thirty-first example in the above-mentioned examples ischaracterized in that the delayed received signals are generated byprocessing the received signals with a filter having a plurality oftime-varying coefficients which alternately take a zero or a non-zerovalue.

The thirty-second example in the above-mentioned examples ischaracterized in that the time-varying coefficients have a zero valueexclusively to each other.

The thirty-third example in the above-mentioned examples ischaracterized in that the time-varying coefficients have a non-zerovalue exclusively to each other.

Above, although the present invention has been particularly describedwith reference to the preferred embodiments and examples thereof, itshould be readily apparent to those of ordinary skill in the art thatthe present invention is not always limited to the above-mentionedembodiment and examples, and changes and modifications in the form anddetails may be made without departing from the spirit and scope of theinvention.

This application is based upon and claims the benefit of priority fromJapanese patent application No. 2008-247273, filed on Sep. 26, 2008, thedisclosure of which is incorporated herein in its entirety by reference.

REFERENCE SIGNS LIST

-   -   1 and 2 received signals    -   3 and 4 loudspeakers    -   5, 6, 7, and 8 echoes    -   9 and 10 microphones    -   11 talker    -   12 and 13 transmission signals    -   14 and 15 mixed signals    -   16 and 17 output signals of a signal processing apparatus    -   18 and 19 digital-analogue converters    -   20 and 21 analogue-digital converters    -   121, 122, 123, and 124 adaptive filters    -   125, 126, 127, and 128 pseudo echoes    -   129 and 130 subtracters    -   300 and 301 delay processing circuits    -   310 and 320 filters    -   330 and 430 clock changing circuits    -   350 and 351 analyzing circuits    -   400 amplitude correcting circuit    -   410 and 420 amplitude processing circuits    -   500 and 501 non-linear processing circuits    -   510, 520, 530, and 540 non-linear amplitude processing circuits    -   511, 531, 3100, 3105, 4100, and 4105 input terminals    -   513 polarity determining circuit    -   514 multiplier    -   515 and 3103 adders    -   516, 536, 3104, and 4104 output terminals    -   600 and 610 frequency analysis synthesizing circuits    -   1000 computer    -   3101 delay element    -   3102 and 4101 coefficient multipliers

The invention claimed is:
 1. A signal processing method of receiving aplurality of received signals, and subtracting echo replicas generatedby a plurality of adaptive filters having said plurality of receivedsignals as an input, respectively, from a plurality of echoes to begenerated from said plurality of received signals, thereby to reducesaid plurality of echoes, comprising: generating delayed receivedsignals by delaying at least one received signal, out of said pluralityof received signals; generating echo replicas by inputting said receivedsignals and said delayed received signals alternately into said adaptivefilters with a period; and controlling said period based upon aperceptual sensitivity to a change in acoustic image localization ofsaid plurality of received signals.
 2. A signal processing methodaccording to claim 1, wherein at least one of said delayed receivedsignals an amplitude-corrected delayed received signal subjected to anamplitude correction.
 3. A signal processing method according to claim1, wherein at least one of said input signals of adaptive filters is anon-linearly processed signal subjected to a non-linear processing.
 4. Asignal processing method according to claim 1, comprising decomposingsaid received signal into a plurality of frequency components, andgenerating the delayed received signals by delaying the received signalfor each of said plurality of frequency components.
 5. A signalprocessing method according to claim 1, wherein said perceptualsensitivity to a change in acoustic image localization is obtained basedupon similarities of the received signals.
 6. A signal processing methodaccording to claim 1, wherein said perceptual sensitivity to a change inacoustic image localization is obtained based upon a power of thereceived signal.
 7. A signal processing method according to claim 1,comprising generating the delayed received signals whose relative delaysto those of the original undelayed received signals take differentvalues and vary with time.
 8. A signal processing method according toclaim 7, wherein said relative delay is an integer multiple of asampling period.
 9. A signal processing method according to claim 1,wherein said delayed received signals are generated by processing thereceived signals with a filter having a plurality of time-varyingcoefficients which alternately take a zero or a non-zero value.
 10. Asignal processing method according to claim 9, wherein said plurality oftime-varying coefficients have a zero value exclusively to each other.11. A signal processing method according to claim 9, wherein saidplurality of time-varying coefficients have a non-zero value exclusivelyto each other.
 12. A signal processing apparatus for receiving aplurality of received signals, and subtracting echo replicas generatedby a plurality of adaptive filters having said plurality of receivedsignals as an input, respectively, from a plurality of echoes to begenerated from said plurality of received signals, thereby to reducesaid plurality of echoes, comprising: a linear processing circuit thatgenerates delayed received signals by delaying at least one receivedsignal, out of said plurality of received signals; an adaptive filterthat generates echo replicas by receiving said received signals and saiddelayed received signals alternately with a period, a plurality ofsubtractors, each of said subtractors that generates echo-reducedsignals by subtracting said echo replicas from a plurality of mixedsignals; and an analyzing circuit that obtains a perceptual sensitivityto a change in acoustic image localization of said plurality of receivedsignals, and wherein said signal processing apparatus controls saidperiod based upon said perceptual sensitivity, and controls coefficientsof said plurality of adaptive filters so that outputs of said pluralityof subtractors are minimized.
 13. A signal processing apparatusaccording to claim 12, comprising an amplitude correcting circuit thatgenerates amplitude-corrected delayed received signals byamplitude-correcting at least one signal, out of said delayed receivedsignals.
 14. A signal processing apparatus according to claim 12,comprising a non-linear processing circuit that generates non-linearlyprocessed signals by non-linearly processing at least one signal, out ofthe signals to be inputted into said plurality of adaptive filters. 15.A signal processing apparatus according to claim 12, comprising: afrequency analyzing circuit that decomposes said received signal into aplurality of frequency components; and a linear processing circuit thatgenerates the delayed received signals by delaying the received signalfor every said plurality of frequency components.
 16. A signalprocessing apparatus according to claim 12, wherein said analyzingcircuit obtains said perceptual sensitivity to a change in acousticimage localization based upon similarities of the received signals. 17.A signal processing apparatus according to claim 12, wherein saidanalyzing circuit obtains said perceptual sensitivity to a change inacoustic image localization based upon a power of the received signal.18. A signal processing apparatus according to claim 12, wherein saidlinear processing circuit performs a processing such that relativedelays of said delayed received signals to said received signals have aplurality of values that vary with a time.
 19. A signal processingapparatus according to claim 18, wherein said linear processing circuitperforms a processing such that said relative delay is an integermultiple of a sampling period.
 20. A signal processing apparatusaccording to claim 12, wherein said linear processing circuit comprisesa filter having a plurality of time-varying coefficients whichalternately take a zero or a non-zero value.
 21. A signal processingapparatus according to claim 20, wherein said plurality of time-varyingcoefficients have a zero value exclusively to each other.
 22. A signalprocessing apparatus according to claim 20, wherein said plurality oftime-varying coefficients have a non-zero value exclusively to eachother.
 23. A non-transitory computer readable storage medium storing asignal processing program for causing a computer to execute a receivingprocess of receiving a plurality of received signals, and an echoreducing process of reducing a plurality of echoes that are generated bysaid plurality of received signals, said signal processing programcomprising: a delayed received signal generating process of generatingdelayed received signals by delaying at least one received signal, outof said plurality of received signals; an echo replica generatingprocess of generating echo replicas by inputting said received signalsand said delayed received signals into said adaptive filters alternatelywith a period; and an echo replica subtracting process of subtractingsaid echo replicas from said plurality of received signals,respectively, wherein said period is controlled based upon a perceptualsensitivity to a change in acoustic image localization of said pluralityof received signals.
 24. A non-transitory computer readable storagemedium storing a signal processing program according to claim 23,wherein at least one of said delayed received signals is anamplitude-corrected delayed received signal subjected to an amplitudecorrection.
 25. A non-transitory computer readable storage mediumstoring a signal processing program according to claim 23, wherein atleast one of said input signals of adaptive filters is a non-linearlyprocessed signal subjected to a non-linear processing.
 26. Anon-transitory computer readable storage medium storing a signalprocessing program according to claim 23, comprising decomposing saidreceived signal into a plurality of frequency components and generatingthe delayed received signals by delaying the received signal for each ofsaid plurality of frequency components.
 27. A non-transitory computerreadable storage medium storing a signal processing program according toclaim 23, wherein said perceptual sensitivity to a change in acousticimage localization is obtained based upon similarities of the receivedsignals.
 28. A non-transitory computer readable storage medium storing asignal processing program according to claim 23, wherein said perceptualsensitivity to a change in acoustic image localization is obtained basedupon a power of the received signal.
 29. A non-transitory computerreadable storage medium storing a signal processing program according toclaim 23, wherein said delayed received signals are generated so thatrelative delays of said delayed received signals to said received delayshave a plurality of values that vary with a time.
 30. A non-transitorycomputer readable storage medium storing a signal processing programaccording to claim 29, wherein said relative delay is an integermultiple of a sampling period.
 31. A non-transitory computer readablestorage medium storing a signal processing program according to claim23, wherein said delayed received signals are generated by processingthe received signals with a filter having a plurality of time-varyingcoefficients which alternately take a zero or a non-zero value.
 32. Anon-transitory computer readable storage medium storing a signalprocessing program according to claim 31, wherein said plurality oftime-varying coefficients have a zero value exclusively to each other.33. A non-transitory computer readable storage medium storing a signalprocessing program according to claim 31, wherein said plurality oftime-varying coefficients have a non-zero value exclusively to eachother.